An emitter-switching circuit configuration comprises a cascode connection of a bipolar transistor having a high breakdown voltage, and a low voltage power MOSFET transistor. Such a configuration is schematically shown in FIG. 1 and is indicated with reference numeral 1. The emitter-switching configuration 1 comprises a bipolar transistor T1 and a MOS transistor M1 cascode-connected together between first and second voltage references, particularly the supply voltage Vcc and ground GND.
The emitter-switching configuration 1 provides that the bipolar transistor T1 is of the HV (High Voltage) type, i.e., a high breakdown voltage transistor, while the MOS transistor M1 is of the LV (Low Voltage) type, i.e. a low breakdown voltage transistor. The bipolar transistor T1 has a collector terminal connected to the supply voltage reference Vcc via an inductive load L1, and a control or base terminal connected to a driving circuit 2. The MOS transistor M1 has a control or gate terminal connected to the driving circuit 2.
The driving circuit 2 comprises a first resistive element RB connected to the base terminal of the bipolar transistor T1, and a Zener diode DZ connected to ground GND. A second resistive element RG is connected to the gate terminal of the MOS transistor M1, and to ground GND via a voltage pulse generator G1. An electrolytic capacitor CB is connected in parallel with the Zener diode DZ, and has across its terminals a voltage equal to VB.
In particular, the electrolytic capacitor CB has the function of storing energy during the bipolar transistor T1 turn-off so that it can be reused during a following turn-on and conduction step of the transistor itself. This is while the Zener diode prevents the base voltage value of the bipolar transistor T1 from exceeding a predetermined threshold.
The emitter-switching is particularly interesting at the present time due to the marketing of bipolar transistors having a square RBSOA (Reverse Biased Safe Operating Area) with a current near the peak current. It also has a voltage equal to the breakdown voltage BVCES between the collector and emitter terminals when the base terminal is short-circuited with the emitter terminal [Breakdown Voltage Collector-Emitter Short], as well as of MOS power transistors having a very low drain-source resistance value in conduction conditions, RDSON and being thus almost similar to ideal switches.
The main advantages of an emitter-switching configuration are an extremely low in-conduction voltage drop (typical of bipolar transistors.) and a high turn-off speed, as readily known by those skilled in the art. When turning off, the current output from the bipolar transistor base terminal is equal to the collector terminal current of this transistor, i.e., a very high current. This causes a drastic reduction both of the storage time and of the fall time, allowing the emitter-switching configuration to operate even at frequencies of 150 kHz.
The driving performed through the driving circuit 2 is very useful and effective in all cases in which the current in the emitter-switching configuration 1 is zero, or very small with respect to the nominal current in the turn-on step.
However, for this driving to be effective, it is necessary that the base current value in the turn-off step IBOFF, multiplied by the storage time tstorage, is near the base current value in the conduction step IBON, multiplied by the turn-on time ton. In other words:IBOFF*tstorage≈IBON*tONThis condition usually occurs when operating at relatively high frequencies and with not too high currents, or better yet, when the bipolar transistor gain value hfe is not too low.
In this case, the driving energy required for the conduction is slightly higher than the energy recovered during the turn-off. It is thus sufficient to supply the base terminal with a very low power to replace inevitable losses.
FIG. 2 shows the trend of voltage VGS values between the gate and source terminals of the MOS transistor M1, of the voltage between the bipolar transistor collector terminal and the MOS transistor source terminal VCS, and of the base and collector currents of the bipolar transistor T1 with reference to a flyback converter operating at a frequency of 100 kHz and having a zero turn-on current since the converter operates discontinuously.
When operating with applications where the current value on the emitter-switching configuration 1 in the turn-on step is not zero, and at frequencies higher than around sixty kHz, the phenomenon of the dynamic VCESAT voltage between the bipolar transistor collector and emitter terminals involves excessive power dissipations. This phenomenon is based upon when the emitter-switching configuration turns on, and there is a certain delay before reaching the static VCESAT voltage value. It is thus necessary to overflow with carriers the bipolar transistor base region as fast as possible to make the VCESAT voltage value decrease and reach the steady value as soon as possible.
It is evident that the higher the bipolar transistor operating frequency, then the more relevant is this phenomenon. The need for a high turn-on base current is in contrast with the need for a reasonable saturation in the turn-off step. In fact, the benefits of an improved turn-on current-voltage crossing would be lost in the turn-off step.
For this reason the need arises for a particular modulation of the bipolar transistor base current in the emitter-switching configuration, which allows both switching steps (turn-on and turn-off) to be optimized and the minimum VCESAT voltage value to be reached.